Electric wave amplification



March 30, 1943. H. W; BODE ELECTRIC WAVE vAMPLIFICA'I'ION Filed Dec.28,1940 3 Sheets-Sheet 1 F/GJ o o w o mwz wo INVENTOP HW 5005 I000[0.000 I 00,000 I.OO0.000

FREQUENCY March 30, 1943. H; w. BODE ELECTRIC WAVE AMPLIFIGATION 3Sheets-Sheet 2 Filed Dec. 28, 1940 FIG. 6

TOTAL NVENTOE H. w 8005 BY ATTORNEY Mafch 30,1943. H. w. BODE 2,315,040

ELECTRIC WAVE AMPLIFICATION Filed Dec. 28, 1940 3 Sheets-Sheet 3 SERIESon m [CA mane Fla It? I00 1,000 10,600 00,000 1,000,000 |o,00o,ooo

- lNl/ENTOR H. W 8005 8) A 7'7'0 NEV v reference will be made inPatented Mar. 30, 1943 ELECTRIC WAVE AMPLIFICATION V Hendrik W. Bode,New York, N. Y., assignor to Bell Telephone Laboratories, Incorporated,New york, N. Y., a corporation of New York Application December 28,1940,Serial-No. 372,073

- 14 Claims.

This invention relates to electric wave ampli fication and moreparticularly to stabilized feedback amplifiers adapted for theamplification of signals occupying a wide frequency range.

A principal object of the invention is to improve the performance andsimplify the construction of stabilized negative feedback amplifiers.Another object is to increase the amount of gain stabilizing feedbackthat may be obtained in an amplifier without sacrifice of otherperformance characteristics.

Another object is to reduce the effects in a feedback amplifier ofspurious or parasitic im-- pedances such as stray capacitancesassociated with amplifier input and output transformers or like couplingdevices.

A further object is to facilitate control of the terminal impedances ofan amplifier as modified by feedback.

Still other objects are to combine in one amplifier the respectiveprincipal advantages of different types of feedback, to effect a smoothtransition over the frequency range from one type of feedback to anotherand to maintain the vectorial product #43 constant over the signalingfrequency range.

The foregoing objects and various'others that will appear hereinafterare achieved in stabilized negative feedback amplifiers employingconcurrent f cedbacks of respectively different types, such as series,shunt, cathode and bridge types. The nature of the present invention andits various features, objects and advantages will appear more fully froma consideration of the following description of the typical embodimentsillustrated in the accompanying drawings.

In the drawings:

Figs. 1 and 2 are schematic showings of feed- Fig. 12 illustrates amodification of Figs. and 11.

Inasmuch as preferred embodiments of the invention chosen for purposesof exposition herein involve cathode feedback, or normal series feedbackor both, it maybe helpful to consider briefly the nature andcharacteristics of these two types of feedback. Fig. 1 illustratesschematically an amplifier having feedback of the normal series typecomprising three amplifying stages I, 2, 3

with input and output transformers 4 and 5, respectively. Although theremay optionally be provided feedback local to the several amplifyingstages, the principal feedback in this type of amplifier is of thevoltage-voltage type and the forward or mu circuit is maintained atsubstantially ground potential, at least in the input and output stages.Thus the anode circuit for the third stage is completed to the groundedcathode thereof through a coupling impedance Z1, and the resultantvoltage drop through the latter is applied in series in the inputcircuit of the first stage. I

A desirable feature of the normal series feedback amplifier asillustrated in Fig. l is that the tend to be especially larg if theamplifier is back amplifiers of the normal series and cathode X ventioncombining series and cathode feedback;

Fig. 4 is a curve diagram to which reference will be made in thedescription of Fig. 3;

' Fig. 5 is a more detailed showing of an amplifier in accordance withFig. 3;

Figs. 6 and 7 show schematically an amplifier combining series and shuntfeedback;

Figs. 8 and 9 comprise curve diagrams to which the description of Figs.6 and 7;

Fig. 10 shows schematically an amplifier combining bridge and shuntfeedback;

Fig. 11 illustrates schematically an amplifier combining bridge andseries feedback; and

adapted for television signals (which mayrange in frequency from cyclesper second to several megacycles. for example), for the transformers maytake the form of the rather bulky coupling networks disclosed forexample in my applicaas disclosed fully in my U. S. Patent 2,123,178,

July 12, 1938.

Negative feedback of the cathode type is exemplified in Fig. 2 which maybe considered as showing an amplifier essentially the same as .that ofFig. 1 excepting for the change in the principal feedback circuit. Thelatter in this case is also of the voltage-voltage type mm the importantdifference that the respective low The shunting capacitances potentialends of the secondary windingvof input transformer 4 and the primarywinding of output transformer 5 are maintained at ground potential. Inthis case. as in Fig. 1, last stage anode current flowing through thecoupling impedance Z: produces a voltage drop across the latter which isapplied in series with the first stage input circuit. The cathodes ofthe first and third stages, it will be noted, are maintained of! groundpotential by the signal voltage drop in the coupling impedance Zn. Thevirtues of the cathode type of feedback appear largely in the asymptoticfrequency range where the fact that the transformer capacitances toground do not appear across the feedback circuit removes the restrictionpresent in the normal series feedback amplifier of Fig. 1.

Another principal advantage of thevcathode type of feedback results fromthe redistribution of input and output tube capacities which it permits.0f the total capacit -associated with the control grid in the inputstage a portion goes to the cathode of the input tube and the remainderto ground. Similarly, the total capacity associated with the anode ofthe output tube is distributed between the output cathode and ground.

- With the normal series type of feedback connection, the cathode andground capacities are indistinguishable, since the cathodes are atground potential. In thecathode feedback circuit, on the other hand, thecapacity between the input grid and ground and the capacity between theoutput anode and ground are removed from the grid-cathode andanode-cathode paths and fall instead across the high impedance or facingto cathode by flowing through the beta circuit.

If the beta circuit includes a direct current resistance the flow ofthese currents produces a dierct current voltage drop across the betacircuit which appears as a grid biasing voltage on the first and thirdtubes. Inasmuch as the beta circuit impedance Z1 should ordinarilyappear as a pure resistance over the low frequency range, the anodecurrent associated with the output stage tends to produce in the firststage an excessive grid biasing voltage drop across the impedance Z1. Toreduce the eflective resistance of the coupling impedance Z1 for directcurrent windings of the input and output transformers.

Now it is shown in my pending application for patent, Serial No.297,069, filed September 29, 1939 (Patent No. 2,242,878, May 20, 1941),that in'a series or cathode feedback system an appropriate ratio must bemaintained between the grid-cathode or anode-cathode capacities and thecapacities acrossthe high impedance windings of the corresponding inputor output trans,- former' ii. a satisfactory asymptotic feedbackcharacteristic is to be obtained. In the normal series feedbackconnection where all of the tube capacities appear in the grid-cathodeor anodecathode'paths, this generally means that large supplementarycapacities must be added across the transformer. The redistribution oftube ca-- pacities in the cathode type feedback circuit on the otherhand permits the desired ratio to be met automatically with the'additionof at most very small supplementary capacities. The reduction of thetotal capacity associated with the. high impedance windings'of thetransformers permits a corresponding increase in the impedance ratiowhich may be obtained with the transformers. In practical cases thetransformer impedanceratios obtainable with the cathode type feedbackconnection ma be as much as two or three times those permissible with anormal series feedback structure and there is a directly proportionalimprovement both in the level of the. input signal delivered-to theamplifier and in the level of'the output power delivered by theamplifier.

The principal difiicultiesencountered in the cathode type feedbackcircuit arise in supplying power to the tubes. The powersupply circuitsthemselves are at ground potential. Since the cathodes 'of the first andthird tubes are off ground potential this means that the screen grid andanode currents of these tubes'must return but not for higher frequenciesone could conceivably shunt the coupling impedance with a largeinductance or choke coil of low resistance. However, the magnitude ofthe inductance theoretically required for this purpose, in view of thefact that as suggested the lower end of the signal range may be 45cycles, produces further complications and is not considered desirable.In a television amplifier especially the beta circuit phase angle is ofsuch importance that the required inductance is very great. The

inductance, moreover, must have negligible distributed capacity to avoiddifficulties at high frequencies.

Still another problem is presented in the cathode type of feedback inview of the fact that the screen grid currents include, in addi tion toa direct current component, an alternating component of signal frequencyplus distortion products. In typical tubes, the signal and distortioncomponents in the screen grid currents are roughly similar to those inthe corresponding anode currents but their level is about 14 decibelslower. If these screen grid current components produce correspondingvoltage drops' in the feedback impedance Z1 the practical benefits ofnegative feedback with respect to gain stabilization and distortionreduction can be realized only to the extent of approximatelyl-i-decibels. The usual remedy of biasing the screen gridthrough aresistancecondenser filter fails when the lower edge of the signal bandis so low as to require a bulky filter condenser havinglarge-distributed capacitance to ground.

In the embodiment of the invention illustrated in Fig. 3, normal seriesfeedback and cathode feedback are intimately and effectively conibined.-The cathodes of the first and third stages are joined together andconnected through a common impedance network Z: to ground. The lowpotential ends of the second ary and primary windings respectively ofthe transformers 4 and 5 are joined together and connected to groundthrough a common impedance network Z4. The two impedance networks Z3 andZ4 may be considered as constituting together a single impedance networkZ5 with an internal connection to ground. Preferably network Z3 is soproportioned that it has substantially constant resistance over onefrequency range and is effective in providing negative feedback in thatrange, and low or substantially zero impedance in a lower frequencyrange. Conversely, impedance network Z4 is preferably so proportioned asto have constant resistance over a low frequency range and low orsubstantially zero impedance over a higher frequency -range.

For high frequencies, such as those of interest with respect to theasymptotic characteristic of zero and-the transformers are thereforeeffectively maintained at ground potential. It will be seen thereforethat in the high frequency range the amplifier is essentially of thecathode feedback type and all of the advantages incithe critical highfrequency range. At low frequencies such that Z3 may be disregarded, itwill be seen that the several cathodes are reduced to substantiallyground potentialand the amplifier becomes essentially of the normal.

series type with its attendant advantages at low frequency in respect tothe supply of power and biasing voltages. It may benoted that in so faras the Fig. 3. amplifier is of the series feedback type, the straycapacitances to ground associated with the anode and grid electrodes ofthe amplifying tubes do notenter into or affect the-external gain of theamplifier inasmuch as they are in the mu circuit. It may be noted toothat there are no high frequency asymptotic requirements on impedanceZ4. Still another significant point that may be mentioned is that thetransformer capacitances to ground and the intere'lectrode capacitancesto the several cathodes are separated, and therefore the possibility isopened for separate treatment and compensation for the capacitances.

Examining now the internal construction of impedance networks Z3 and Z4,an arrangement is shown whereby a smooth transition from series tocathode feedback is effected. More particularly, impedance network Z3 isrepresented as comprising an inductance L and a shunting resistance R1,and Z4 is represented as comprising a capacitance C shunted by aresistor R2. In actual practice the two impedance networks may be muchmore complex than here indicated, hence the simple networks shown shouldbe considered as representing the equivalent circuits of the actualnetworks. The equivalent circuits then are preferably so proportionedthat L/C is equal to Him. If more particularly R1 and R2 are equal, thenit can be shown that the impedance Z of the'combination is equivalent toa resistance Rat all frequencies.

In other words the feedback coupling impedance is independent offrequency despite the transition from one type'of feedback to another.The solid lines in Fig. 4 show in curve diagram form the manner ofvariation of the three impedances with respect to frequency. Since Fig.3 has been indicated as showing simplified equivalent ,circuits for Z;and Z4, it may be well to point out that the equivalency need not holdat frequencies far removed from the transition fre quency range. Thatis, at very low frequencies it is not essential that inductance L retainits character as such or that R1 remain a resistance or of unchangedmagnitude so long as the over-all impedance Z3 is negligible. Similarlyfor. Z4, it is not essential that its components maintain the characterindicated at frequencies where Z4 is substantially zero. The dottedlines in Fig. 4 apply to a case where a certain amount of cathodefeedback is retained atlow frequencies, as for example where aresistance effective at low frequencies is interposed in series with theinductance L in the Z3 structure of Fig. 3. Again it will be observedthat the transition is smooth and that the total feedback is constantthroughout the signal frequency range.

With regard to the cross-over frequency, that is, the frequencyrepresented by the intersection dent to that type of feedback areretained in I of curves Z3 and Z4 in Fig. 4, it is desirable in the caseof the television amplifier assumed that it be a comparatively lowfrequency of the order, of a few kilocycles. On the one hand thecrossover frequency should be high enough that alternating currentcomponents in the screen grid lead are effectively by-passed to thecathode. An upper limit on the cross-over frequency is fixed by thetransformer capacitances to ground, for these establish a minimum valuefor the capacitance element C in Fig. 3, understanding that thesecapacitances are a part of the capacitance C there represented. Whereasthis last consideration might fix an upper limit of a few hundredkilocycles for the cross-over frequency a lower frequency may be founddesirable, particularly if a non-uniform gain characteristic, such asone increasing with frequency from say 50 kilocycles upward, is to beincorporated in the amplifier.

In this discussion it has been supposed that the total beta circuitimpedance, and therefore also the total amplifier gain, should be flatwith frequency. The provision of a variable gain characteristic,however, is also feasible. Thus, a shaping impedance network dominant athigh frequencies may be interposed in series with one of the terminalleads of Z2 and another dominant at low frequencies may besimilarlyassoshould be understood too that the provision of a betacircuit impedance that is not fiat with frequency is not inconsistentwith s being constant throughout the signaling frequency range, for as Bvaries, a may be made to vary in inverse" relation by shaping theinterstagenetworks in well-known ways.

Fig. 5 shows in greater detail an amplifier substantially conformingwith Fig. 3 and specifically adapted for amplification of televisionsignals occupying the frequency range from 45 cycles to 3 megacycles,and further adapted for compensating the non-uniform attenuation of atransmission line repeater section. The amplifier tubes I, 2 and 3. arepentodes and are connected in tandem by suitable coupling impedances. Inthe first stage the screen grid is provided with a condenser by-pass toground and the suppressor grid is directly grounded. In the second stagea resistance and shunting condenser are provided in the cathode lead toprovide control grid bias and local feedback. The screen grid isprovided with a condenser by-pass to ground and the suppressor grid istied directly, to the cathode as it is in tube 3 also. In the thirdstage the screen grid is provided with a condenser by-pass to thecathode and it is connected to the biasing battery through a resistance.

The beta circuit network Z3 comprises an inductance it of 10millihenries and a resistor 12 of 70 ohms connected in series across aresistor ll of 333 ohms. The value of resistor i2 is such .as to providean IR drop suitable for biasing the transiormer-to-ground capacitance is0.0014 microfarad. Condenser i4 is shunted by a resistor IS in serieswith the anode and screen grid voltage source, the latter beingrepresented by the battery symbol and assumed to have negligibleinternal impedance. Where resistor I2 is not employed resistor l may be383 ohms; the same value as resistor II. If a resistor 12 is employed itis compensated by shunting'resistor IS with the inverse of resistor l2,which in this case is 1560 ohms, or the resistor l5 may be used aloneand assigned an equivalent value, namely, 275 ohms.

A grid leak connection for the first stage grid is provided by a.resistor Il of 0.16 megohm, for example, and if desired aresistance-condenser combination It may be interposed in seriestherewith as shown and the several values so proportioned asadditionally to.provide low frequency 48 control.

Th circuit elements of networks Z3 and Z4 that are shown in Fig. 5 butnot described above are employed for shaping the beta circuittransmission characteristic at frequencies well above the cross-overfrequency to compensate for the rising attenuation frequencycharacteristic of a preceding repeater section of a coaxial conductortransmission system. These elements have no substantial effect atfrequencies in the vicinity of the cross-over region but only atfrequencies in the range above say 50 kilocycles.

The cross-over frequency in the Fig. 5 circuit as above described isapproximately 5 kilocycles or'about one octave below the mean frequencyof the signal band. This frequency is high enough that excessivefeedback of the output screen grid current components is avoided.

In another embodiment of the present invention series feedback and shuntfeedback are advantageously combined, in a manner to be described withreference to Fig. 6. The amplifier represented in this figure is thesame as the series feedback amplifier illustrated in Fig. 1, (the seriesfeedback coupling impedance Z1 being now designated 26,) excepting forthe addition of a shunt feedback connection which may extend, forexample, as shown from the last stage anode to the first stage controlgrid and which includes an tions being the same, that could be achievedwitha single type of feedback. Another special object is to facilitatecontrol of the input and output impedances of an amplifier as they aremade to appear by feedback action. Still another object relates tocompensation of spurious impedances associated with the coupling networkof a series feedback type of amplifier.

Bearing in mind that the voltage fed back to the input grid in a seriesor shunt feedback amplifier is ordinarily a small fraction of the totaloutput voltage, it will be understood that the series feedback impedanceZ6 is small compared with R0, the impedance presented by the highimpedance windings of the input and output transformers. Likewise theshunt feedback impedance Z1 is large compared with Re. In a typical caseZ6 may be Bil/100 and Z7 may be 100R). Accordingly it may be supposed toa fair degree of approximation that these ratios are so extreme that R0in series with Z6, or R0 in parallel with Z1, is substantiallyequivalent to R0 alone.

The anode current I in the last stage produces a voltage drop IRo acrossthe output transformer and a voltage drop 12: across impedance Zn. Thesecond of these voltage drops is the series feedback voltage. The shuntfeedback path comprises a potentiometer made up of impedance Z1 and theR0 of the input transformer. With the assumptionthat Z7 is much largerthanR-o, the potentiometer causes a fraction Ro/ Z1 of the voltageacross the output transformer to be fed back by this path. The totalfeedback voltage E is therefore where Y-: is the reciprocal of Z: and Ztis the transfer impedance from the output stage anode Reciprocally. ifthe maximum obtainable shunt feedback is known the required seriesfeedback impedance Ze to bring the total feedback to a desired greatervalue can be-ascertained.

If, as indicated schematically in Fig. 7, the

series feedback coupling impedance comprises aresistance K in parallelwith another impedance Z, particularly simple relations are obtained forthe case of constant total p or feedback. In this case.

- l KZ 1 K wT=( -m we 1m 4) The necessary shunt feedback impedance Z7 isthen In other words, the required shunt feedback impedance is aresistance of a certain magnitude in series with an impedance that is amultiple, integral or non-integral, of impedance Z.

The impedance Z shunting the resistance K'in Fig. '7 may actuallycorrespond, for example, to an unavoidable parasitic or spuriousimpedance such, for example, as the capacitance to ground of the inputand output transformers. The effect of such capacitance in a seriesfeedback amplifier has been considered hereinbefore and it has beenshown that the stray capacitance may reduce the feedback practically tozero before the top of thesignal band is reached. To compensate for thespurious capacitance a shunt feedback circuit is provided as in Fig. '7in which the feedback impedance comprises a capacitance and resistancein series with each other and of such respective magnitude as to satisfyEquation 5. The division of feedback between the two feedback paths,shunt and series, is shown diagrammatically in Fig. 8 for a typicalcase.

If the parallel impedance Z is more complicated, a more complex divisionof the frequency spectrum between the two types of feedback is obtained.For example, if the impedance Z in Fig. 7 comprises in addition to thetransformer capacitance a parallel connected inductance element orbranch, the feedback division might appear as in Fig. 9, the totalfeedback remaining constant'over the frequency spectrum. .The inductivebranch assumed may correspond, for example, to an anode current sourcein series with a choke coil.

Although in the examples described with reference to Fig. 7 theimpedance Z has been assumed to represent a parasitic element, it mayalso be regarded as an impedance branch added deliberately to thecircuit to secure some desirable result. For example, it is wellknownthat the apparent or active input and/or output impedance of a feedbackamplifier is not necessarily the same as its passive impedance butrather a function of the feedback. In a series feedback amplifier theactive impedance Za is much larger than the passive impedance Z while ina shunt feedback amplifier the converse is true. More specifically, forseries feedback,

' a value for Z which is of the .proper'order of magnitude and whichvaries in the desired manner with frequency an active amplifierimpedance can be established whichfollows any prescribed course withrespect to frequency at any magnitude between the very high impedanceproduced by pure series feedback and the very low impedance produced bypure shunt feedback.

Since the series feedback is large and the shunt feedback small whenimpedance Z is large and conversely for small values of Z, it is evidentthat the active impedance Za must be large when Z is large and smallwhen Z is small. The exact relation for large values of p is quitesimple and can be shown to be 1 In other words the active impedance ismerely a multiple of whatever impedance Z is inserted.

Whereas Equation 8 assumes constant #1 any desired active impedance Z5can be obtainedfor any given feedback characteristic, represented by Zt,by designing the shunt and series feedback impedances in accordance withthe following relations:

lying the present invention two examples involving bridge type feedbackwill be described with reference to Figs. 10 to 12.- In the Fig. 10amplifler the input and output circuits comprise resistance bridgesproviding bridge type feedback through a beta circuit path that includesa conventional constant resistance equalizer containmg the disposableimpedance'branches Zn and Z21. A shunt feedback path comprising theimpedance Z1 is provided as in Fig. 6. The equalmet in the bridgefeedback circuit is introduced to permit the feedback through the bridgepath to be controlled as a function of frequency. The same control canbe had by transferring the impedance elements Z21 and Z11 to the inputcircult bridge as indicated in Fig. 12. For a constant resistanceequalizer the voltage E1 fed back to the input control grid through thebridge feedback circuit is where the denominator represents thefrequency variation introduced by. the equalizer and K1 represents theconstant losses ofthe two bridges.

For extreme impedance levels, that is, for very large values of Z1, thefeedback through the shunt path is i E7=K2Y7 (12) where K: is a constantdetermined by the bridge impedances. If a constant total feedback K1 isdesired, then v the t is, a resistance in series with a multiple of Z21.The active impedance presented by the Fig. 10 amplifier resembles aconstant resistance,

corresponding to the impedance resulting from bridge feedback, inparallel with an impedance Z21. Over a frequency range where Z21 islarge portion to the following relation:

A o n Z6 o 2. R0 1i (14) that is, as a multiple of Zn in parallel withRe. In this case the amplifier impedance will be inverse to thatobtained in th case of Fig. 10 and it will vary approximately as theseries combination of Zn and the constant resistance.

W th respect to the combinations illustrated in Figs. 10 to 12, it isnoted that the asymptotic impedance Zc should vary with frequency'inprocharacteristics of a bridge type feedback circuit are generally poorbecause of high losses in the input and output bridges which make itdifiicult to obtain larg values of feedback in the signal band. Thecombination disclosed permits the bridge feedback to be maintained overat least part of the bandbut still permits the eventual feedback tooccur in an asymptotically superior path. These combinations may also befound useful in obtaining amplifier impedances d partplify'signalcurrents occupying a frequency range I of at least several octaves, saidamplifier comprising a plurality of amplifying stages and a plurality ofgain stabilizing feedback circuits embracing said'stages, saidfeedbackcircuits providing feedback of respectively different types, and thefeedback frequencycharacteristics of said proportioned that the totalfeedback provided by said feedback circuits is substantially constantover a, wide frequency range.

circuits being so proportioned that the several types of feedback arepreponderant in respectively different portions of the frequencyspectrum and the total feedback provided by said 8. A broad bandamplifier comprising a plurality of amplifying stages and input andoutput transformers, a stabilizing feedback circuit of thevoltage-voltagetype embracing the proximate windings of saidtransformers and comprising a feedback coupling impedance, said couplingimpedance comprising at least two series-connected sections grounded attheir Junction, and said sections having such respectiveimpedancefrequency characteristics that the feedback is of the normalseries type in one frequency range and of the cathode type in anotherfrequency junction, said sections having respective impedfeedbackcircuits is substantially constant over the frequency range occupied bysaid signals.

' 3. An electric wave amplifier comprising an odd number of amplifyingstages, each of said stages-comprising a space discharge amplifyingdevice having a plurality of electrodes including a cathode, a feedbackcircuit of the cathode feedback type embracing said stages andcomprising feedback coupling means providing cathode-toground waveimpedance that is substantially greater in the first and last of saidstages than in an intermediate stage, and a feedback circuit of thenormal series feedback type embracing said stages.

4. An amplifier comprising a plurality of amplifying stages, afeedbackcircuit of the cathode feedback type embracing said stages and afeedback circuit of the normal series feedback type embracing saidstages, said cathode feedback predominating at high frequencies and saidseries feedback predominating at lower frequencies.

5. A multistage amplifierhaving amplifier input and output transformers,a feedback circuit of the normal seriestype comprising saidtransformers, and a feedback circuitof the cathode type comprising saidtransformers,.said feedback circuit of the normal series type comprisinga feedback couplingimpedance that has negligible impedance in theasymptotic frequency range of said amplifier. I i

6; A multistage amplifier comprising amplifier input and outputtransformers, a voltage-voltage feedback circuitcomprising saidtransformers and a feedback coupling impedance across said feedbackcircuit, said coupling impedance being grounded at a. point electricallyintermediate its terminals.

7. An electric wave amplifier, a negative feedback circuit of one typefor said amplifier com-.

prising a feedback coupling impedance, said coupling impedancecomprising two series-connected sections having differentimpedance-frequency characteristics, a second negative feedback circultof another type for said amplifier comprising at least one of saidsections, said sections being so ance-frequency characteristics suchthat in a transition frequency range the impedances of said sectionsvary in opposite senses with respect tofrequency, the series impedanceof said two,

sections being constant throughout said transition frequency range.

10. An amplifier adapted for the direct amplification of televisionsignals or the like comprising space discharge amplifying devices in aplurality of stages, a cathode feedback circuit and a series feedbackcircuit embracing said stages, a feedback coupling impedance common tosaid circuits and grounded at a point electrically between itsterminals, and a space discharge current sourc connected in said seriesfeedback circult.

11. In an electric wav amplifier, the method of controlling the apparentimpedance of said amplifier which comprises controllably feeding backwaves amplified therein to the input of said amplifler in such relationas to tend to increase the apparent impedance of said amplifier,concurrently feeding back waves amplified therein to th said input insuch relation as to tend to decrease the apparent impedance ofisaidamplifier, and so adjusting the relative preponderance of the twofeedbacks indifferent portions of the frequencyrange that the saidapparent input impedance varies in a predetermined manner over thefrequency range.

12. An electric wave amplifier comprising a. plurality of amplifyingstages and feedback circuits of the series and shunt types respectivelyembracing said stages, said.- feedback circuits being so proportioned asto be principally effective in respectively different portions of thefrequency spectrum.

13. An amplifier inaccordance with claim 12 in which the total 3 of saidfeedback circuits combined is constant over the frequency range of thewaves being amplified.

